G11 Block Diagram
G11 Power Supply
A pair of thyristors in a controlled full-wave bridge configuration provide a 100Hz rectified supply. This is smoothed by an active smoothing circuit to give a DC supply of 1.56V. The thyristors only conduct for a short period when they are triggered in the latter half of each mains half-cycle. Stabilisation of the DC output voltage against both mains voltage changes and load variations is achieved by controlling the conduction angle of the thyristors.
The trigger pulses are produced by a silicon controlled switch with a phase control circuit which senses the mains input and the DC output voltages. At switch-on, the phase control circuit provides slow start-up. In addition to the normal CRT beam current limiter in the signal circuits, further protection is provided by reducing the power supply output voltage if the EHT current becomes excessive. A monostable gating circuit inhibits spurious thyristor triggering by spikes which may be present on the incoming mains supply. Over-voltage protection is incorporated.
Two diodes A/B of the four-diode encapsulation D4005 along with the two thyristors SCR4018/20 form a controlled full-wave (100Hz) bridge rectifier circuit. Thyristor conduction, and therefore the power into the load is controlled by changing the phase of the thyristor trigger pulses, and by this means a stabilised DC output is obtained. All four diodes of D4005 form a second full-wave bridge rectifier to provide a separate 100Hz supply for the trigger pulse generator SCS4061.
The input bridge circuit is such that the receiver chassis is
floating at about half-mains potential irrespective of which way round
the mains connection is made. Two fuses FS1301/02 are fitted, so that,
regardless of which mains input lead is “live”, there is protection for a
short circuit between chassis and true earth during servicing.
For reliability reasons, the mains input to the two thyristors is taken via paralleled plugs and sockets. VDR1307 in conjunction with the RF choke L1305 and C1304/06 suppresses incoming mains interference.
The choke and capacitors are also partly responsible for preventing RF interference generated by the thyristors from entering the mains supply. In addition to this mains filter, the remaining interference resulting from the very rapid changes of peak current through the thyristors is eliminated by the two-chokes-on-one-core assembly L4009A/B. This choke is extremely important in order to avoid interference to MW and LW radio receivers.
Trigger Pulse Generator
This is powered by the 100Hz unsmoothed rectified mains waveform from D4005. The waveform is attenuated and delayed by the phase shift network R4044/C4058, and the resultant rising voltage across C4058 is applied to the anode of the silicon controlled switch SCS4061. The SCS gate is fed from the same source via the attenuator R4059/60; C4062 is not large enough protection purposes. An output pulse appears at the SCS cathode after the anode voltage has risen to meet the falling gate voltage. This is the source of trigger pulses for the two thyristors. The phase relationship of the anode and gate waveforms is such that a trigger pulse can only be produced during the back half of the positive input half-cycles. C4064 is for flashover protection. The timing of the trigger pulses is controlled by T4045 and then passed via the pulse inhibit gate T4072 and the trigger pulse amplifier T4068/77 to the thyristors.
Trigger Pulse Phase Control
T4045 is placed in parallel with the trigger pulse generator charging
capacitor C4058. The conduction of T4045 is deter-mined by the
rectified mains supply via R4059/53, and also by the DC stabilised
output voltage fed back from C4029 via R4024/41 /42 etc. The controlled
conduction of T4045 restricts the charge entering C4058, thereby
determining the amplitude of the SCS anode waveform. In this way the
transistor controls the phase of the trigger pulses which, in turn, vary
the conduction periods of the thyristors, and thus effectively
stabilises the voltage across C4029 against both changes in HT load
current and mains input voltage variations. C4039 prevents the 100Hz
ripple in the yet unsmoothed DC output from being fed to 14045.
Adjustment of the stabilised DC output voltage is provided by R4042 in
the bias network for the control transistor.
Provision is made for remote control “standby” facility, and under this condition the emitter of T4045 is grounded. T4045 now “bottoms” and prevents the formation of any trigger pulses for the thyristors. There is therefore no HT output from the power supply.
The Darlington Pair T4032/33 is initially cut off during the start-up period because the emitter of T4032 rises quickly in potential via R4031, but the base of T4033 can rise only as fast as C4034 can charge via R4030/36. D4066 prevents excessive reverse V” of T4032/33 during this period, and afterwards becomes non-conductive. After a short time, normal working conditions prevail and the load current is approximately equally divided between R4031 and T4032. The bias for the smoothing transistors is obtained by the potentiometer R4030/35 between the input and output voltages. Because the input voltage is stabilised (approx. 175V) it follows that the output is also (156V). C4034 is the smoothing capacitor which determines the ripple on the output voltage. C4040 provides no further smoothing at 100Hz, but gives a low impedance path for the AC components of the HT load current.
Short-circuit protection is incorporated in order to protect T4032/33 from failure in the event of the HT becoming short-circuited. In the short interval between the occurrence of a short-circuit and the resulting rupture of the 1A fuse FS4037, the potential on C4034 remains substantially constant, and both transistors bottom at a safe current decided by the voltage drop across R4052. The excessive HT current causes a volts drop across the protection resistor R4036, and the voltage across C4029 falls somewhat, but the situation is not basically altered in the short remaining time until FS4037 ruptures which then relieves T4032/33 of all stress.
Normally, D4021 is non-conductive, but conducts in the event of an excessive overload or partial short-circuit of the HT supply. Under this condition, although the HT output falls, the zener diode maintains the forward bias on T4033 in order to ensure that T4032 continues to pass sufficient current to rupture FS4037. R4019 limits the diode current to a safe value.
After the 1A fuse has blown the thyristors now operate with no load current but the voltage across the reservoir capacitor C4029 hardly changes. The voltage does not rise because the automatic voltage stabilising circuit alters the firing point of the thyristors to a later phase position. When under this condition the receiver is switched off or disconnected from the mains, the reservoir capacitor (400p) voltage cannot run down quickly and remains charged to about 175V for a long period. This feature should be remembered when taking the backplate off a receiver during servicing.
R4067 is an isolated fuse resistor driven by the glow switch 4038. The local heating effect produced by current through the resistor is capable of melting a fusible solder link, which in turn, breaks the HT feed from the thyristors. Normally the glow switch is inoperative but if the HT voltage at C4029 rises above about 195/205V the glow switch ignites (ionises) and heats its internal bi-metal contacts which then close. Current then passes through R4067 causing it to warm up. The glow is now extinguished because there is no longer any voltage across the electrodes. When the contacts cool and open again, the process repeats unless in the meantime the over-voltage has disappeared. Thus a short duration HT rise has no lasting effect, but a persisting excessive voltage eventually opens the fuse link associated’ with R4067. The higher the over-voltage the more rapidly is R4067 heated and the fuse opened. Depending on the degree of over-voltage, a period of between 20 to 80 secs may elapse before the spring fuse associated with R4067 ruptures. The fuse may be reset by the use of a soldering iron and 60/40 (no other grade) tin/lead alloy solder.
CRT Beam Current Protection
T4085/86 are normally non-conductive. However, under certain fault conditions e.g. a short circuited video output transistor, the EHT current taken by the picture tube loads the EHT circuit and the line output stage in a dangerous manner. The operation of the normal beam limiter is not designed to deal with an overload of this magnitude. The excessive current through the EHT overwind causes the voltage across C3123 (in the line time-base) to swing negative to chassis thus turning on T4085 via R3121. This in turn causes the current amplifier T4086 to conduct and drag down the emitter voltage of the power supply control transistor T4045. The effect can be so great as to reduce the power supply output voltage to virtually zero. This results in the CRT heater (which is fed from the line time-base) cooling down thus reducing the overload on the EHT supply. The power supply voltage is now allowed to return, and over a period of a few seconds the cycle repeats until the fault is cleared. D4090 provides EHT flashover protection for T4085.
At switch-on, a large surge of mains current flow through both halves
of the dual-PTC thermistor R1303A/B. The current flowing through
section “A” also passes through the degaussing coils L1801/02. The whole
thermistor rapidly becomes very hot and both halves increase in
resistance causing the degaussing current to fall to an insignificant
value. The idling current now flowing through section “B” keeps the
component hot. The two halves are thermally coupled, and the heat
transfer from section “B” to section “A” keeps the resistance of “A”
high, thereby holding off the degaussing current.
After switching off and the thyristor has cooled down, the degaussing circuit is then ready for another switch-on cycle. To prevent beam landing errors caused by currents induced in the degaussing coils by the line deflection field, the coils are short-circuited at line frequencies by C1803.
TDA2590 Line Oscillator Combination
The TDA2590 IC contains all the circuitry necessary to produce an accurately phased line drive signal (pin 3) for the line output stage, together with field sync (pin 8), and a burst gating pulse (pin 7).
The sync separator is driven by the incoming luminance signal at pin
9. The input networks R2035/C2043/R2033 etc. improve synchronisation
under adverse signal conditions. Field sync pulses derived from the
composite sync train are available at pin 8.
The line oscillator produces a triangular waveform across the external timing capacitor C2021. The line flywheel time constant C2009/19/23 and the associated resistors, is normally long but is shortened automatically by the internal time constant switch which operates during VCR operation and whenever the line oscillator loses synchronisation. The VCR switch feeding pin 11 closes when the VCR button is depressed on the programme selector unit.
The trigger pulse generator provides a suitable drive waveform at pin
3 for the line timebase. The mark/space ratio of this wave¬form is
controlled internally to suit the operating conditions of the line
output stage; R2025 allows a manual adjustment to centre the control
range of the automatic circuitry.
A burst gating pulse output at pin 7 is superimposed on a line blanking pedestal, providing a so called “Sandcastle Pulse”. However, the blanking component is not used in this receiver.
Normally, the IC is powered (pins 1 and 2) from the 12V LT rail via D2015. However, at switch-on the LT rail is at zero volts, and the IC is initially powered from the 156V HT rail via R2010, D2015 being non-conductive at the moment. This allows the line output stage to function sufficiently well to partially power the 12V rail. D2015 now conducts, the line output drive builds up, and eventually the LT feed becomes the full 12V. Very little power now reaches the IC via the self-start resistor R2010.
The output waveform from the line oscillator combination IC is amplified by T3102. To obtain sufficient power this transistor is fed from the HT1 rail (156V). R2018/C2027/L3111 prevent VHF interference radiation. The step-down transformer L3107 provides a low impedance switching waveform for the line output transistor T1401. R3104/C3105 damp the primary of L3107. R3108/09 damp any ringing of the secondary, preventing the possibility of the line output transistor being turned on during flyback.
Line Output Stage
T1401 functions as the usual fast bi-directional switch, the
transistor passing current in both directions, but in this case the
transistor itself only partially provides the conventional “efficiency
diode” energy recovery action. In the main this is effected by the E-W
diode modulator. R3108 limits the peak base current of the transistor,
and C1402 protects its base/emitter junction against EHT flashover
transients which may appear on the collector and be transferred to the
base via the collector/base self-capacitance The high working voltage
capacitor C3131 tunes the line output transformer.The line output
transformer L3157 is physically small because it mainly operates merely
as a choke, and the deflection coils L1804 are directly capacitively
coupled to the Line output transistor.
The HT supply voltage feeding the line transformer primary L3137F/G is also stored across C3135/36 in series. When the output transistor turns on and is “bottomed”, this voltage is thus applied both across the primary and the deflection coil network. The resultant approximately exponential build-up of current and the energy recovery action which follows, provides sawtooth currents through the transformer and the deflection coils. The higher frequency elements of the line currents are decoupled by C3129 from the HT rail.C3135/36 together provide “S” correction and DC blocking from the output transistor to the scan coils. Line linearity is corrected by L1501 in series with the scan coils. L1501 is a non- linear saturable reactor biased by a permanent magnet so that it has a higher inductance (and therefore a high loss) at the start of stein than at the end.
Optimum linearity on the left hand side of the picture is achieved by rotating the magnet. The coil is damped by R1502 to prevent ringing.The HT4 supply for the CRT G2 controls (approx. 800V) is obtained from D3115/6 which rectify the 1.2kV line flyback pulse on the collector of the line output transistor. R3117/C3114 are for limiting and smoothing. To obtain a suitable control range the earth return for the G2 potentiometers is taken to the 156V rail via R1618. The CRT guns may be turned off by means of removing their G2 voltage with the switches U1617D, E and F.Winding “D” on the line output transformer provides a 23Vpp negative pulse waveform of 6.3V RMS for the picture tube heaters. Windings “F” and “G” drive the diode modulator D3132/33. Winding “H” in parallel with “F” and “G” in series provide coupling between the two limbs of the line transformer core in order to minimise the leakage inductance between windings. Winding “B” and the scan rectifier D3138 provide the LT3 and LT4 lines (37V). Winding “A” and the scan rectifier D3147 provide the LT1 line (16V) to the 12V stabiliser IC5073. The potential divider R3145/46 provides negative line flyback pulses for the vision detector unit U5600. Winding “E”, in conjunction with the 156V line and the flyback rectifier D3139, provides the HT3 (165V) line to the RGB output stages. Winding “C” provides positive line flyback pulses for the Line Oscillator Combination IC2510, line blanking and the Colour Demodulator IC6530. Winding “J L N Et Q” together constitute the EHT overwind, and the 5kV focus voltage is taken off part way along the chain. The resistor values in the thick-film focus unit U3138 are chosen to give a focusing voltage range of approximately 3.8 to 5.1kV.The line output stage incorporates a high-level diode modulator circuit which provides East-West pincushion and Keystone corrections of the raster by modulating the line scanning current with a combination of parabolic and sawtooth field-rate waveforms.
The modulator also provides adjustable picture width by controlling the scan current through the line deflection coils.In comparison with the deflection field for the 110′ delta-gun picture tube, the pronounced pincushion shape of the 20AX horizontal deflection field necessitates the application of about twice as much E-W correction; and in addition the S-correction must be modulated. North-South correction is not required. The resulting increased power requirements of the diode modulator is compensated for by the reduced loading on the line output stage due to the absence of high-current-consuming dynamic convergence circuits. D3133 is a specially developed heavy current device (BY223) able to withstand the high flyback voltage present on the collector of the line output transistor.
E-W pincushion distortion is corrected, without modulating the EHT and ancillary services, by the E-W diode modulator driven by the E-W drive circuit T2150 etc. L3137 basically forms part of a “lossy” bridge-coil circuit in series with the line deflection coils, and thereby forms a means of the modulator being able to change the width of the raster. In practice the bridge-coil consists of an isolation transformer in order to accommodate the linearity and S-correction circuits.
Due to the nature of the 20AX deflection field, more S-correction is required for the centre horizontal line than for the top and bottom lines. The depth of modulation applied to the S-correction component must therefore be different to that applied to the deflection current. The deflection coil current flows through the series S-correction capacitors C3135/36, but the correction introduced by C3136 is suitably modulated by the action of the diode modulator bridge-coil L3137.
Because the S-corrected scanning current is field-rate modulated, further correction is necessary to ensure that the line linearity correction is optimum for both the centre horizontal line and the top and bottom lines. This is provided by the tap on the linearity control L1501, which allows the bridge-coil modulation current to pass through part of the winding.
Because the bridge-coil reactance forms part of the total line output transformer flyback tuning, during modulation when the line-rate voltage across the bridge coil changes, this would undesirably introduce a change in the tuning at a corresponding rate. This change is nullified by sympathetically varying the line transformer tuning via C3128.
In the unmodulated condition, C3128 with the bridge coil, forms part of a balanced bridge, thereby rendering C3128 ineffective. During modulation, as the voltage across the bridge coil becomes smaller, the line output tuning tends to rise. However, the reduction in the voltage across the bridge coil has unbalanced the bridge, and a capacitive AC current now flows through C3128. This electronically variable capacitance is reflected across the line transformer primary thus preventing the resonant frequency rising.
D3132/33, which are across the line output transistor, pass the majority current of the flyback energy, and therefore, although the transistor conducts (in reverse mode) during this period also, it plays only a small part in the energy reclaim action. D3132 alto conducts during the scan and as a result of the rectifying action generates a mean positive voltage which is used to power the E-W driver stage. The current drawn from this supply determines the modulation depth of the diode modulator.
Pincushion Correction and Width Control
To correct for the pincushion distortion concavity at the sides of the raster, it is necessary to modulate the line scanning current parabolically at field-frequency. The E-W Diode Modulator in the line timebase performs this function, and is driven by a low impedance field parabola waveform from the E-W driver circuit.
The field sawtooth voltage across the deflection coil current sensing resistor(s) R2104/05 is converted to parabolic form by the Miller Integrator stage T2119. The network R2115/D2116/17/ R2123 rounds the ,top and bottom of the sawtooth applied to 12119, and thereby adjusts the shape of the ends of the parabola, and hence also, of the four corners of the raster. The shape of the basic parabola at the collector of T2119 is modified by the net¬works C2112/R2113/C2111 and R2134/C2122/R2121, The amplitude of the output parabola is adjustable by R2137, the low end of which is taken to a suitable DC bias point to avoid interaction with the width adjustment.
T2140/49/50 form a Darlington Trio driver stage having a high current gain and a low output impedance. The emitter voltage for T2150 is supplied by the diode modulator, and R2147 provides negative voltage feedback for the Darlington. A small amount of the input sawtooth waveform is added via R2114 into the output to tilt the parabola. The choke L3134 ensures that the E-W drive circuit presents a high impedance at line frequency to the diode modulator; the choke is ineffective at field frequency. At the same time the choke along with the by-pass capacitor C2151 prevents any line frequency components from reaching T2150. R2133 adjusts the DC level of the generated parabola, and achieves its width control function via the action of the diode modulator.
Diode-split Method of EHT Generation
Instead of employing a conventional voltage trebler, ENT is obtained from a diode-split EHT winding on the line transformer. The encapsulated EHT overwind consists of four separate similar single layers, the interlayer capacitances of which are used as reservoir capacitors for three of the four EHT rectifier diodes. The diodes are an integral part of the transformer and eliminates the need for a separate tripler unit. In order to obtain an EHT output of 25kV, each single layer provides a peak flyback voltage of between 6 to 7kV. Due to the action of the rectifiers and reservoir capacitors the AC voltage between adjacent layers is virtually zero and therefore there are practically no dielectric I osses.
The screened EHT lead forms a reservoir capacitor for the fourth diode, while R3159 and the capacitance of the CRT aquadag coating are used for smoothing. This screened lead and R/C network also prevents unwanted line radiation. In the event of EHT flashover, R3159 limits the current flow in order to protect the line output transistor. The EHT current drawn by the picture tube produces a negative DC voltage across R3122/C3123, and this is used to operate the beam current limiter as required
Line and Field Blanking
The positive going field flyback pulse from the field output filter coil L2092A and C2099 etc. is fed to the clipper diode D2158 via the auto bias network C2156/57. This diode passes the tip of the waveform only on to the monostable T2159/64 The pulse triggers the monostable where T2159 is normally off”‘ and T2164 “on”. The cross coupling components are C2163 and R2163. The critical leak away time constant C2162/R2162 controls the width of the output field blanking pulses at the collector of T2164, which are made as wide as possible (approx. 22 lines, 1.4mS) consistent with the field retrace period, in order to blank test and teletext signals which are present during the field flyback. D2166 is to protect the base/emitter junction of T2164 when the base is driven below chassis potential.
R6037/38 mix line flyback pulses from the line transformer and field blanking pulses from T2164 to form a composite blanking signal for the luminance/chrominance module U6200 in which they are further amplified and limited to a common level.
A 50Hz sawtooth is generated within the IC, and appears at pin 10. It is shaped and amplitude adjusted by R2052 and R2058 respectively. The frequency is determined by the timing network R2048, C2053/56 and the setting of the hold control R2045. Synchronisation is effected by sync pulses fed into pin 13.The sawtooth waveform is applied via C2080 to pin 7 of the IC, together with a voltage from the deflection coil current sensing resistor(s) R2104/05 to provide overall feedback. The combined signal is amplified in the IC and passed to a pulse-width modulator. An internally generated 150kHz triangle waveform is also fed into the modulator the action of which is to produce a train of pulses. The width of these pulses is dependent on the instantaneous amplitude of the sawtooth. The pulse-width modulated waveform output at pin 3 is passed via C2072 and pin 2 to the output stage of the IC. This comprises a “Totem Pole” pair of power output transistors which are alternately switched between cut-off and bottomed states.
The output from pin 16 of the IC is demodulated by the low-pass filter L2092/C2093. The resulting 50Hz waveform, from which the RF has been removed, is fed via C2099/2100 to the deflection coils L1804G/H. Clamp diodes D2094/95 and D2087/ 89 are used to prevent voltage overshoots at pin 16. Overshoot at the’ earthy” end of L2092 is restrained by the damping network R2106/C2107. C2070. ferrox bead 2085, R2073/ C2074, C2088/96, C2050/68 and C2069 are fitted in order to remove the higher order components of the switching frequency. A variable shift control is unnecessary with the “20AX” system, but R2102 which “bleeds” a small current past the output coupling capacitor(s) C2099/2100, provides a small upward displacement of the raster. To prevent crosstalk from thd power output switching, the earlier stages in the IC are isolated from chassis by the protection diodes D2062/63. The varying load current during the field scanning cycle is decoupled by R2067/ C2097/98 etc. from the line-scan rectified LT supply rail; the small remaining disturbance is eliminated by feeding an effectively equal but inverted signal into the system via R2108.
The field synchronising pulse is fed from circuitry which is remote from the field timebase IC. To avoid possible failure of the IC in the event of EHT flashover, the stopper resistor(s) R2037/44 is fitted.
CONVERGENCE CORRECTION AND PURITY
(A) Static Correction, Purity and N-S Symmetry
The object of static correction (convergence) is to cause the landing positions of the two outside electron beams (red and blue) to coincide with the central (green) beam in the centre of the picture tube screen. The static correction unit is combined with the purity adjuster and a N-S raster shape adjuster. The combination consists of four pairs of magnetic rings, each pair coupled by pinion gears, and is fitted on the tube neck behind the deflection coils. The rings generate multiple-pole fields which suitably shift all three or just two of the beams as required.
(B) Dynamic Correction
Convergence errors in the outer areas of the display are corrected by four additional multiple-pole fields. Two of these are produced by a four-pole unit wound on the deflection coils ferroxcube ring and driven by line and field currents. The other two multi-pole fields are obtained by means of balancing currents through each half of the line and field deflection coils.
The drive for the field convergence controls is derived from the field scan current.
The scan current passing through R1525 develops a sawtooth voltage which provides the drive to the symmetry controls R1531/39 and the four-pole unit L1804. The field scan current flows in reverse directions during the top and bottom halves of the screen. Diodes D1535/36 conduct during the first and second halves respectively of the field scan in order that the current through the four-pole unit shall be in the same direction, rather than reversed, during both the top and bottom halves of the screen. The resultant multi-pole field provides similar horizontal movements of the two outer beams (red and blue) at the top and bottom of the screen.
The four diodes D1526/27/31/32 and the associated resistors form a differential circuit across the two halves of the field deflection coils. By unbalancing the scan current flowing in the two halves of the coils, a multi-pole field is created which shifts the two outer beams (red and blue) in a vertical direction. Due to the switching action of the diodes whereby D1526/31 and D1527/32 conduct only during the first and second halves of scan respectively, R1528 affects the top of the screen only and R1529 the bottom.
The drive for the line convergence is derived from the line scan current.
Correction of errors on the centre horizontal axis (red/blue crossover) is obtained by the difference current provided by the differential coil 11516. The resultant multi-pole field shifts the two outer beams (red and blue) in a vertical direction along the centre horizontal axis.
The resistive circuit associated with PC3 is in series with the line scan current. The resultant voltage across the resistive element (R1506/15 etc) causes a difference current to flow through the two halves of the line deflection coils. The multi-pole field which is created in the coils again corrects vertical errors of the two outside beams with respect to the centre beam on the centre horizontal axis. The correction current is clamped by D1509 in order to prevent the dynamic adjustments from affecting the static convergence. The adjustment of PC3 has effectively three positions, but for convenience a standard five pin socket is used for the link; position “B” is the same as “C”. Opposite polarity corrections are provided at positions “A” and “0”.
The line frequency current for driving the four-pole unit L1804 is derived from the line scan current passing through the primary of L1504. Depending on the position of its core, positive or negative line pulses are present at the secondary of the coil, and these cause a current to flow in the four-pole windings. The resulting multi-pole field corrects errors of the two outside beams (red and blue) in a horizontal direction across the centre axis.
The G11 Tuner
Tuner Unit and IF Amplifier
The low noise UHF “PIN” tuner used in this receiver gives improved signal handling properties compared with earlier designs. Gain control is effected by an internal PIN diode attenuator controlled by varying the currents in two PIN diodes via the AGC system. The PIN attenuator behaves virtually as a linear UHF attenuator. The gain control current flows out of the tuner via R5007 which limits the current at maximum gain to 9mA. The TCA270 forms a sink for this current (as it also does for the IF control current). Forward gain control is used for the IF amplifier. The “takeover” characteristic is modified by the action of the zener diode D5012 which suitably “catches” the IF gain control voltage. A feature of the AGC system is that a disturbance is injected into the AGC loop by the field sync pulse train. This results in a “kink” in the video waveform during the field blanking period, and the effect is minimised by the choice of the second-order time constant R5615/C5620.
The IF channel consists of a medium gain, choke-capacitor-coupled, three stage amplifier with little selectivity. The main selectivity, hole-traps etc., is provided by the selectivity filter between the tuner and the IF strip. The IF coil in the tuner is part of the selectivity filter. Although screened, the untuned IF amplifier is liable to direct pick up of high field strength CH1 405 line VHF television signals. The bridge-T filters L5612/C5613 and L5621 /C5612 etc., reject these sound and vision carriers if they are present when the receiver is located in the immediate vicinity of a CH1 television tower (i.e. Crystal Palace).
IF signals are applied to the IC to one side (Pin 2) of a differen-tial input. The other side (Pin 1) being decoupled by C5617. The IC contains separate synchronous demodulators for video and AFC. Tuning for video is by L5626/C5625 and for AFC by L5630/C5629. Coupling for the AFC quadrature coil is by stray capacitance (Cx) between the IC pins. Also contained in the IC are a video amplifier, a peak-acting (sync tip) AGC detector and AGC amplifier. The AGC system is line gated and exercises control by sinking current from the tuner and IF bias chains. D5012 catches the IF gain control to improve the take over control characteristic. The take over point is adjustable by R5013. This control has little effect at maximum gain when both control pins (A El. 51 are “sinking” maximum ctirrent.
Depending on the tuning state pin 11 (AFC control O.P.) will either
source or sink current to/from the tuner AFC circuit. The potential
divider R1912/16/17 across the 33V stabiliser (TAA550) sets the
operating centre point to 6 volts. The AFC feed from the
TC.A270 is discriminated whenever the tuning draw is open and also when a selector button is depressed.
T5060 base is supplied with field sync pulses. These are almost
removed by the integrator R5062/C5061, and under normal operating
conditions T5060 is non-conductive. Once
BIAS 6 MHz SOUND locked in, the AFC characteristic is excellent but from an initial start or during channel changing the AFC tends to lock out due to the luminance carrier being severely attenuated by the 41.5 MHz adjacent sound hole-filter. Under this condition excess AFC control current is being sourced, and the off-tune video and sync circuits will be swamped by receiver noise. The DC level of the input signal to T5060 (field syncs having bsen replaced by noise) now rises, C5061 charges, and T5060 bottoms. R5059 is thus placed across the AFC detector output, and drains excess current. As the tuner local oscillator approaches the correct tuning point, the selected channel is suddenly pulled in by the AFC operation; T5060 then becomes non-conductive again. The action of T5060 also prevents the tendency for the AFC to capture the 33.5 MHz sound signal when the aerial input is temporarily interrupted.
L5641 /58 and their associated components are hole-filters to remove
the unwanted sound and chrominance signals from the chroma and luminance
6 MHz intercarrier sound selectivity is provided by the top capacity coupled bandpass pair L5637/38 which also incorporates a broad band 4.43 MHz notch filter in the primary. R5057/R5561/65 match the I K11 luminance delay line L5056.
Luminance/Chrominance Control Unit
R6234/C6232 give HF lift to the luminance signal. R6230 provides the required level of current drive from the low impedance emitter follower T5064 into the luminance amplifier within the TBA560 IC. R6231 decides the black-level current of the amplifier.
The input “sandcastle” waveform at connection 7 is used to operate the luminance black-level clamp, and also to gate out the burst signal. There is a diode characteristic on pin 10 of the IC, and the circuit only responds during the tip of the input pulse. To ensure that black-level clamping is not partially influenced by changes of sync amplitude (via the sandcastle pulse), the front portion of the sandcastle pulse is removed during the line sync period. This operation is effected by T6005 which is driven into a “bottomed” state by the positive line sync pulses in the composite luminance signal applied to its base. R6006/C6007 provide auto-bias and limiting.
The line and field blanking signals at connection 6 are combined by R6037/38. After limiting to a constant level by T6258, the resulting composite blanking signal is applied to the luminance and chrominance amplifiers within IC6229. This blanks the luminance signal to “blacker-than-black” during the retrace periods of the line and field timebases, and also removes the burst from the signal passing through the 2nd chroma amplifier.
R6228 is the load resistor for the internal emitter follower amplifier feeding pin 5 of the IC. During blanking, this amplifier is cut off, and R6221 sets up a blanking level of 1.5V at this point.
The high-pass filter C6226/R6233, along with the push-pull flatly tuned coil L6227/R6233, form the chroma selectivity circuit. The first chroma amplifier is gain controlled by the ACC voltage from IC6520 (TBA540); C6249/R6245 is the anti-hunting network. The network R6236/37/38/40, sets the DC working point of the burst amplifier with the IC.
The output burst signal from the gated burst amplifier is used to lock the regenerated reference subcarrier. In order that locking occurs on a phase of B-Y, the burst signal is shifted in phase (approx. 90° lag) by the adjustable network L6039/C6050/ 86051.
The control commands for variation of brilliance, contrast and saturation (colour), are in the form of decoupled DC potentials potted down from the 12V line. For any given setting of saturation, the chroma signal amplitude will track with adjustments of the preset contrast control. Also, the two signals track together during normal operation of the beam current limiter function.
The saturation control potential is also applied to a three-position pluggable flylead PC2. In the “normal” position, when the colour-killer operates it shunts this control potential to chassis, thus preventing the 2nd, chroma amplifier from working. In the “kill” position, the chroma channel is inoperative bceause there can be no control potential. With the flylead unplugged, the “enable” position, the chroma channel remains active (enabled), irrespective of whether the colour-killer is operative or not.
C6243 is the charge storage reference for the black-level clamp. The operating point of the black-level clamp is controlled by the brightness controls R1704/R6212, and also by the overriding action of the beam limiter control voltage fed into connection 2. As long as the CRT mean beam current is below 1.5mA, D6211 remains cut off, and the brightness control functions normally. Above 1.5mA, however, the negative going control voltage forward biases the diode, thus reducing the brightness as required. The operation of the beam limiter is entirely automatic; no manual adjustment is provided. In the event of serious overload, the maximum excursion of operation is limited by the forward conduction of D6011, thus protecting the IC Er C6001. This diode also provides protection in the event of EHT flash-over. Major EHT current overloads due to possible fault conditions, are dealt with by an additional beam current limiter (see power supply).
Subcarrier Reference Combination
A 4.43 MHz crystal is externally connected between pins 1 and 15 to form the regenerated subcarrier reference oscillator. The burst signal is synchronously demodulated by the reference, to produce a DC potential for the reactance control. The latter effectively magnifies the capacitance value of the trimmer C6045, and completes the phase locked loop. The oscillator reference voltage generated on a phase of B-Y is inverted by the centre tapped coil connected between pins 4 and 6. After being phase shifted 90° by C6036/R6042 to form the R-Y subcarrier reference, the B-Y and R-Y outputs are passed to IC6530. The H/2 component from the demodulated burst signal is compared with the H/2 square-wave from the PAL bistable signal fed into pin 8. The resultant signal is used to supply the ACC detector.
With no input burst signal the potential on pin 9 is set by the ACC level preset R6058 to 4V. When burst is applied, and if the PAL bistable-switch phase in the TCA800 is correct, this potential will drop until ACC control is established at about 1V. If however, the bistable phase is incorrect, the potential on pin 9 will momentarily rise to about 10V. This is passed on to the TCA800 (IC6530), and inhibits the bistable action. As a result of the removal of the switching waveform from pin 8, the potential on pin 9 drops, thus allowing the bistable to start up again. Having “missed a beat” the PAL bistable switch phase should now be correct and the ACC control potential be 1V. The colour-killer which operates in the absence of a burst signal, or during incorrect identification, is activated as the potential on pin 9 rises above about 2.5V.
The 4.43 MHz U and V chrome signals to be demodulated are fed into pins 10 and 11 via DC blocking capacitors C6022/23. Pins 13 and 15 are the B-Y and R-Y subcarrier reference axis inputs for switching the U and V demodulators. The V demodulation axis is commutated during alternate lines by the PAL switch.
After filtering and de-weighting, the B-Y and R-Y video outputs of the two demodulators are matrixed to generate G-Y, and then these three signals are maxtrixed with the EEB, ER and EG camera signals.
The input line pulse via R6027 at pin 8 is shaped into a narrower window pulse which is used to trigger the PAL bistable and to gate the feedback clamp potentials of the RGB output waveforms. The PAL bistable drives the PAL switch, and its H/2 square-wave output is provided at pin 12. This signal is AC coupled to the TBA540 for polarity comparison. The bistable is inhibited for identification purposes when the input DC voltage at pin 14 rises above about 6.5V. This provides a sufficiently high noise immunity margin above the zero colour burst level of 4V, to eliminate spurious identifying.
The three 2.2 LiF clamp capacitors at pins 2, 4 and 6 arc charged via comparator circuits gated by the internally formed window pulse. The capacitors store the blanking potentials derived from each of the blanked RGB outputs at pins 3, 5 and 7. These potentials are compared with a single internal stable reference potential to effect the clamping of the RGB output signals. In
this way any common mode and differential drifts between blanking levels of the RGB output potentials are minimal. Avoidance of differential drift is most important in order to avoid unwanted colouration of the lowlights.
The RGB output signals have low impedance, and are sufficiently large ( >5Vpp), to directly drive simple output video transistors. These are mounted on a common heat sink, and have a relatively low stage gain of about 20-25 times. Therefore, during operation, the contribution that the output stages themselves make towards thermal DC drifts are relatively small. At maximum drive, the black-to-white amplitude available exceeds 100V. Adjustable DC-level controls are no longer fitted in the RGB circuits; the grey scale lowlights are set up solely by the G2 potentiometers.
To avoid unwanted variation of black level during grey-scale
highlight blue and green drive adjustments, constant black-level gain
controls are used. Taking the green channel as an example, the current
through R6087 raises the DC black-level potential at the RHS of R6094 to
the similar potential as that existing at the LHS. Thus adjustment of
the green drive has minimal effect on the black-level and lowlights.
R6088 ensures the correct emitter bias for T6093 during its minimum
current excursion. C6084/ 98/110 in conjunction with their relative
parallel resistors provide H F lift. The capacitor values are different
to ensure that each of the three output stages have the same bandwidth.
EHT flashover protection is provided by internal diodes which, in conjunction with external series resistors R1608 etc., limit dissipation within the IC should flashover occur.
Amplifier/Limiter TBA750 (105510) and Sound Output
The 6 MHz intercarrier sound signal is amplified, amplitude limited and demodulated in the TBA750 IC. The limited 6 MHz signal, via C5023/27, generates a quadrature carrier reference signal for the balanced FM demodulator used in the IC. The volume is controlled by the current obtained from the volume control R1705. D1706/07 improve the control characteristic at the low volume end of the control (slider near the bottom).
The 2.5W sound output stage operates under “class A” condition and is DC stabilised by the negative feedback path R5035/44 which embraces the audio amplifier within the IC. Fixed tone compensation takes place via the negative feedback network C5033/39 etc. from the output stage into pin 16 of the IC. Provision is made for a variable tone control to be connected to plug “5F”, in which case the link between pins 3 and 4 will be removed.